Many communication systems up-convert electromagnetic signals from base band to higher frequencies for transmission, and subsequently down-convert those high frequencies back to their original frequency band when they reach the receiver, processes known as up-conversion and down-conversion (or modulation and demodulation) respectively. The original (or base band) signal, may be, for example, data, voice or video. These base band signals may be produced by transducers such as microphones or video cameras, be computer generated, or transferred from an electronic storage device. In general, the high frequencies provide longer range and higher capacity channels than base band signals, and because high frequency radio frequency (RF) signals can propagate through the air, they can be used for wireless transmissions as well as hard-wired or fibre channels.
All of these signals are generally referred to as radio frequency (RF) signals, which are electromagnetic signals; that is, waveforms with electrical and magnetic properties within the electromagnetic spectrum normally associated with radio wave propagation.
A typical artefact resulting from the down-conversion process are blocking signals, also termed spurious components, generated by the radio which can desensitize the receiver. These blocking signals appear at an unwanted radio frequency, and can desensitize the receiver by raising the noise floor, causing a reduction in the overall gain of the receiver, or a combination of both.
Many different receiver architectures have been proposed and used for mitigating the effect of blocking signals. These include super-heterodyne, image rejection, direct conversion, near zero-IF conversion and harmonic mixing architectures. A brief summary of these architectures follows.
The super-heterodyne receiver uses a two-step frequency translation method to convert the signal at RF to a base band signal. First, the incoming signals and corruptive noise are passed through a band pass filter that attenuates out of band signals and passes the desired signal. At this stage some of the blocking signals that are out of band are filtered. The desired signal, plus residual blocking signals, are amplified and mixed with a first local oscillator. This causes both a down-conversion and an up-conversion in the frequency domain. Usually the down-converted portion is retained at the so-called “Intermediate Frequency” (IF). Further filtering is performed on the signal at the IF frequency using a discrete device. This filter is a band pass filter and retains the radio channel required and further reduces the residual blocking signal. The signal is then mixed with a second oscillator that causes frequency translation to base band. The disadvantages of the super-heterodyne architecture include the requirement for an expensive off chip IF filter, a frequency plan fixed in hardware, and locations of spurious signals that are fixed relative to the RF wanted signal in hardware, meaning that they cannot be changed using a software change.
There are several image rejection architectures that have been proposed, and among these, the two most well known are the Hartley Image Rejection Architecture and the Weaver Image Rejection Architecture. Here a spurious signal is created and is located at a fixed location in frequency relative to frequency of the wanted signal. This spurious signal is commonly referred to as the imagining frequency. The imagining blocking signal is removed using a combination of phase shifters and adders that are applied directly to the radio signal itself or/and the local oscillator (LO) signal. Some methods employ poly-phase filters to cancel the image components. Generally, either accurate phase shifters or accurate generation of a quadrature-mixing signal are employed in these architectures to cancel the image frequency. The amount of image (or blocker) cancellation is directly dependent upon the degree of accuracy in producing the phase shift or in producing the quadrature mixing signals. Although the integratability of these architectures is high, their performance is relatively poor due to the required accuracy of the phase shifts and quadrature oscillators. Another disadvantage here is the location of the blocker signal (or image frequency) is fixed relative to the wanted signal and cannot be moved to another location.
Direct conversion architectures perform the RF to base band frequency translation in a single step. The RF signal is mixed with a local oscillator at the carrier frequency. There is therefore no image frequency, and no image components to corrupt the signal. Direct conversion receivers offer high integration, but also have several important problems. Classical direct conversion receivers have thus far proved useful only for signalling formats that do not place appreciable signal energy near DC after conversion to base band. Though direct conversion does not suffer from blocking signals in general, there are several typical problems found in integrated direct conversion receivers. The noise near base band (i.e. 1/f-noise) corrupts the desired signal, the local oscillator leaks, which creates DC offsets and causes desensitization, noise inherent to mixed-signal integrated circuits corrupts the desired signal, and large on-chip capacitors are required to remove unwanted noise and signal energy near DC.
The near zero-IF conversion architecture is similar to the direct conversion architecture, in that the RF band is brought close to base band in a single step. The desired signal is not brought exactly to base-band however, and therefore DC offsets and 1/f noise do not contaminate the signal. Image frequencies (i.e. the blocker) are again a problem as in the super-heterodyne and image rejection architectures. Specific problems encountered with these architectures include a second down conversion being performed in the digital domain due to spurious issues, a fixed frequency of the image based on the frequency planning which cannot be changed, the need for several balanced signal paths for image cancellation, corruption of the desired signal due to noise inherent to mixed-signal integrated circuits, and the filters used to filter the IF signal inherently contributes to the frequency planning, making them standard specific.
The harmonic mixing architecture uses a number of mixing signals that are phase shifted by some desired amount. If x(t) is the incoming RF signal, and α1, α2, and α3 are the mixing signals, the output of a harmonic mixing structure equals x(t)*(α1+α2+α3). In this example, there are assumed to be three mixing signals. Here, α1, α2, and α3 are constructed so that when they add they have significant energy at the wanted carrier frequency. The frequency of α1, α2, and α3 are usually the same. In all cases, aα1+α2+α3 will have other frequency components other than the wanted carrier frequency. This produces a fixed spurious response. The disadvantage here is the spurious components are fixed based on the frequency planning of the additive signals (for example α1, α2, and α3).
The virtual local oscillator (VLO) receiver architecture described in commonly owned U.S. Pat. No. 6,727,764, the contents of which are incorporated herein by reference, is directed to the generation of signals used in the conversion process. The virtual local oscillator receiver architecture has properties that overcome the image-rejection problems associated with heterodyne receivers and transmitters, and the LO-leakage and 1/f noise problems associated with direct conversion receivers and transmitters. FIG. 1 generally illustrates the main concepts of the virtual local oscillator receiver architecture.
The VLO receiver 10 essentially consists of two mixers 12 and 14 that are connected together. Other parts of the receiver are not shown to simplify the schematic. At the LO ports of the mixers 12 and 14 the signals φ1 and φ2 are applied such that the overall RF signal x(t) is multiplied by a signal having significant power at the RF carrier frequency. FIG. 2 illustrates example φ1 and φ2 waveforms that can be applied to VLO receiver 10, and the resulting base band output signal φeff, which represents the actual desired local oscillator frequency. The resulting base band output signal φeff=φ1*φ2 has significant power at the RF frequency, but in practice there will be power generated in places other than the RF carrier frequency.
This power is denoted as unwanted power, the amount which is determined by the timing delay and frequency of signal φ2. FIG. 3 shows a possible φ1*φ2 spectrum where the desired signal at ωrf has good power, but due to timing delay of φ2, additional tones will appear and are placed at harmonics of φ2 away from ωeff. In the present example, the tones appear at ωrf+200 MHz and ωrf−200 MHz. Unfortunately, this unwanted power will down convert signals located at the unwanted power frequencies. For example, if there is unwanted power at ωrf+200 MHz in φ1*φ2 and there is an out off band blocker signal 20 at ωrf+200 MHz as shown in FIG. 4, this blocker 20 will eventually be down converted on top of the desired signal 22 if left uncorrected, as shown in FIG. 5. Since the two signals overlap, filtering will be ineffective for removing the blocker signal 20.
Accordingly, the VLO receiver receiving such an overlapping signal will drop the transmission since the overlapping signals will result in corrupted data that cannot be properly processed or simply resolved by the receiver.
It is, therefore, desirable to provide a method and system for reducing or eliminating blocker signals from an RF input signal that can desensitize a VLO receiver and potentially cause loss of received data.